System and method for providing distributed amplification

ABSTRACT

A distributed amplifier, or if desired, an oscillator can be constructed using an active transmission line. The active transmission line, in turn, can be constructed using an auxiliary conductor inductively coupled to the primary conductor. The auxiliary conductor is driven by the primary conductor through an active shunt network distributed along the transmission line. The auxiliary conductor is placed close enough to the primary conductor so that the two conductors have a substantial amount of mutual inductance as compared to their self-inductance. In a variation of this system, the transmission line can be operated in differential mode. In one embodiment, a combination of conductance and transconductance are used to achieve gain in the transmission line for high frequency signal transmission. With absorbing terminations, the active transmission line constitutes an amplifier. With reflective terminations, the active transmission line constitutes an oscillator.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is related to concurrently filed, co-pending,and commonly assigned U.S. patent application Ser. No. ______, AttorneyDocket No. 10030059-1, entitled “SYSTEM AND METHOD FOR PROVIDING ALOSSLESS AND DISPERSION-FREE TRANSMISSION LINE”; and U.S. patentapplication Ser. No. ______, Attorney Docket No. 10031011-1, entitled“SYSTEM AND METHOD FOR PROVIDING A DELAY LINE AND/OR FINITE IMPULSERESPONSE FILTERS USING A LOSSLESS AND DISPERSION-FREE TRANSMISSIONLINE”, the disclosures of which are hereby incorporated herein byreference.

TECHNICAL FIELD

This invention relates to distributed amplifiers and more specificallyto systems and methods for providing amplification using active feedbackalong a transmission line.

BACKGROUND OF THE INVENTION

The bandwidth of ordinary amplifiers is typically limited by parasiticcapacitances. As frequency increases, the impedance of such capacitancesdecreases, thereby tending to short-circuit the signal to ground or toother circuit nodes. A common approach to extend the bandwidth ofamplifier circuits consists of adding inductors at judicious locations.When properly designed, the combination of such inductors and thesurrounding parasitic capacitances results in an equivalent impedanceretaining a substantial magnitude over a wider frequency range than theparasitic capacitances alone, thereby extending the amplifier bandwidth.Transmission lines constitute a particular example of this technique.Such lines have distributed capacitance to ground tending toshort-circuit the signal, but the existence of distributed inductancealong the line results in an overall impedance which is resistive at anyfrequency (at least in a somewhat idealized theoretical case).

Some amplifier circuits (known as “distributed amplifiers”) takeadvantage of this fact. Typically, they consist of multiple ordinaryamplifier circuits connected in parallel through inductors (see e.g. D.M. Pozar, Microwave engineering, 2^(nd) ed, John Wiley & Sons, 1998).The parasitic capacitances associated with the inputs and outputs of theordinary amplifier circuits combine with the inductors to formtransmission lines. Therefore, these parasitic capacitances do notcontribute to restrict the bandwidth of the ordinary amplifier circuits.They are said to be “absorbed” by the transmission lines. As a result,distributed amplifiers typically achieve a much larger bandwidth thanindividual ordinary amplifier circuits used in isolation. However, thistype of distributed amplifier suffers from some shortcomings. One of theproblems is that the transmission lines connecting the ordinaryamplifier circuits together unavoidably have losses due to the seriesresistance of the inductors. Such losses cause the input signal to waneas it travels along the transmission line connecting the inputs of theordinary amplifier circuits, thereby limiting the number of amplifierswhich can be put in parallel. A second shortcoming is that the overallgain of the distributed amplifier grows only lineary with the number ofelementary amplifiers used in the structure. It would take a largenumber of elementary amplifiers to achieve a large gain.

BRIEF SUMMARY OF THE INVENTION

The limitations of existing distributed amplifiers can be overcome byusing a transmission line including distributed active feedback elementscausing the signal to grow as it travels along the line. One system forachieving gain in a primary conductor is by use of an auxiliaryconductor inductively coupled to the primary conductor. The auxiliaryconductor is driven by the primary conductor through an active shuntnetwork distributed along the transmission line. In a variation of thissystem, two pairs of conductors including a first and second primaryconductor and a first and second auxiliary conductor can be operated indifferential mode. As will be seen later, the distributed active shuntnetwork can be particularly simple in differential mode.

A distributed amplifier can be constructed using an auxiliary conductorinductively coupled to the primary conductor. The auxiliary conductor isdriven by the primary conductor through an active shunt networkdistributed along the transmission line. The auxiliary conductor isplaced close enough to the primary conductor so that the two conductorsare inductively coupled (i.e. have a substantial amount of mutualinductance compared to their self-inductance). In a variation of thissystem, two pairs of conductors including a first and second primaryconductor and a first and second auxiliary conductor can be operated indifferential mode.

In one embodiment, a combination of conductance and transconductance areused to cancel losses and control dispersion in the transmission linefor high frequency signal transmission. The signal is not assumed to bebinary in amplitude, and the transmission line can operate on analog aswell as digital signals. In such an embodiment, transconductance isachieved in a differential transmission line by inducing a signal fromeach transmission line into closely coupled parallel lines, addingactive elements between each of the coupled lines to a common groundplane and influencing the current through each active element by thesignal on the opposite transmission line.

In another embodiment the transmission line provides gain whileremaining dispersion-free. The total gain grows exponentially with linelength and there is no fundamental limit to the length over which thetransmission line will provide gain.

In another embodiment the bi-directional nature of the transmission lineenables the implementation of active resonant line segments for use ason-chip frequency references. Thus, an oscillator can be constructedwithout the use of crystals or other control devices.

In still another embodiment, the transmission line can be used as adelay line, for example, in a finite impulse response (FIR) filter.

The foregoing has outlined rather broadly the features and technicaladvantages of the present invention in order that the detaileddescription of the invention that follows may be better understood.Additional features and advantages of the invention will be describedhereinafter which form the subject of the claims of the invention. Itshould be appreciated that the conception and specific embodimentdisclosed may be readily utilized as a basis for modifying or designingother structures for carrying out the same purposes of the presentinvention. It should also be realized that such equivalent constructionsdo not depart from the invention as set forth in the appended claims.The novel features which are believed to be characteristic of theinvention, both as to its organization and method of operation, togetherwith advantages will be better understood from the following descriptionwhen considered in connection with the accompanying figures. It is to beexpressly understood, however, that each of the figures is provided forthe purpose of illustration and description only and is not intended asa definition of the limits of the present invention.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a schematic representation of one embodiment;

FIG. 2 shows a single slice of the structure of FIG. 1;

FIGS. 3A, 3B, 3C and 3D depict embodiments of lumped implementations ofdifferential and non-differential circuits;

FIG. 4 shows a termination network;

FIGS. 5, 6, and 7 show graphs of a test system;

FIG. 8 shows one modification to achieve network gain;

FIG. 9 shows a generalized shunt network;

FIG. 10 shows a representative alternative circuit;

FIG. 11 is an approximation of a transmission line by a ladder networkof inductors and capacitors;

FIGS. 12A and 12B are illustrated circuit architectures implementing afinite impulse response filter; and

FIG. 13 is an equivalent circuit model of an active coupled line.

DETAILED DESCRIPTION

Before beginning the description, it should be noted that envisionedapplications include transmission of critical high-frequency a.c.signals within large chips, or over long distance transmission lineswith the concepts taught herein being used in repeaters to boost andcontrol signal dispersion. In addition, accurate delay lines, on-chiposcillators and frequency references, high-speed output drivers anddistributed electrostatic discharge (ESD) protection structures, finiteimpulse response filter and other circuit elements could also bedesigned around the concepts discussed herein. Amplifier array chipsbased on these concepts could be inserted in series with longprinted-circuit board (PCB) traces in order to split their length andthereby boost the bandwidth of such traces.

FIG. 1 shows a schematic representation of one embodiment, in which asymmetrical distributed structure is represented. The distributedstructure shown in FIG. 1 includes differential transmission lines suchthat transmission line 11 ⁺ carries the exact opposite signal fromtransmission line 11 ⁻. Coupled to, but not electrically connected with,each transmission line is an auxiliary conductor, such as conductor 12 ⁺and conductor 12 ⁻. Conductor 13 is the common return path ground. Thecross-section structure is essentially invariant along the direction ofthe x axis. Each conductor 12 ⁺ and 12 ⁻ is connected to common ground13 by a number of conductance and transconductance elements spaced alongthe transmission line corresponding to points 11 a ⁺ to 11 n ⁺ onconductor 11 ⁺ and points 11 a ⁻ to 11 n ⁻ on conductor 11 ⁻. In theembodiment shown, transconductance is achieved by controlling a currentdevice, such as current device 14 aGm⁺ from the differential “opposite”transmission line. Thus, the transconductance elements associated withtransmission line 11 ⁺ are controlled by the signals at the respectivepoint on opposite transmission line 11 ⁻. In FIG. 1, x is the directionof a.c. transmission and, as will be discussed, can be bi-directional.

For better clarity, FIG. 2 shows a schematic of a single slice 20 of thestructure shown in FIG. 1. Conductors 11 and 12 (for convenience, thenotation conductor 11 means both conductors 11 ⁺ and 11 ⁻ and similarly,conductor 12 means conductors 12 ⁺ and 12 ⁻) are close enough to eachother such that the capacitance between them is not much smaller than isthe capacitance between conductors 11 and 13 or between conductors 12and 13. The spacing between the positive and the negative side of thedifferential transmission pair is not critical. Optimally, the positiveand negative conductors should be far apart enough so that thecapacitance between them is smaller than the capacitance between themand conductor 13. A factor of 3 smaller would be ample. For example, thespacing of the conductors should be such that the capacitance betweenconductors 11 ⁺ and 11 ⁻ is less than 60 pF/m while, as shown in Table2, the capacitance (C10′) between conductors 11 and 13 (ground) is, forexample, 173 pF/m. The capacitance between conductors 12 ⁺ and 12 ⁻ isless than 7 pF/m, while the capacitance (C20′) between conductors 12 and13 is, for example, 21.2 pF/m. These examples assume the othercapacitances have the values shown in Table 2, discussed hereinafter.The other parameters are as shown in Table 2.

Ideally, active shunt network 21 would be truly distributed along thelength of the transmission line. In this case, conductance 14 aG⁺ wouldbe a made of continuous resistive material, whereas 14 aGm⁺ would be asingle, very wide transistor. However, in common integrated circuittechnologies, it would be difficult to cross-connect the controlelectrodes between the two sides of the differential structure in atruly distributed implementation. This is so because lines 14 a ⁺ and 14a ⁻ (as well as lines 14 b ⁺ and 14 b ⁻; 14 c ⁺ and 14 c ⁻, etc.) eachwould be continuous and thus physically unable to cross. Instead, a goodapproximation of the distributed shunt network can be obtained by lumpedshunt circuits placed at regular intervals along the transmission lines,as shown in FIG. 1.

Each pair of coupled transmission lines is characterized by a set ofcapacitances and inductances per unit length, as well as the seriesresistance per unit length of each conductor. These parameters arelisted and described in Table 1. As will be discussed, wave propagationin conductors 11 ⁺ and 11 ⁻ is lossless and dispersion-free indifferential mode if the shunt element values are chosen as follows (theprime mark is used to indicate per unit length): $\begin{matrix}{{G2}^{\prime} = \frac{{R1}^{\prime} \cdot {C12}^{\prime}}{{M12}^{\prime}}} & (1) \\{{Gm2}^{\prime} = {{R1}^{\prime}\frac{{C10}^{\prime} + {C12}^{\prime}}{{M12}^{\prime}}}} & (2)\end{matrix}$ TABLE 1 Transmission line parameters Parameter DescriptionC10′ capacitance per unit length between conductors 11 and 13 C20′capacitance per unit length between conductors 12 and 13 C12′capacitance per unit length between conductors 11 and 12 L1′self-inductance per unit length of conductor 11 L2′ self-inductance perunit length of conductor 12 M12′ mutual inductance per unit lengthbetween conductors 11 and 12 R1′ series resistance per unit length ofconductor 11 R2′ series resistance per unit length of conductor 12

Lossless and dispersion-free propagation is available over the wholefrequency range where the above conditions can be satisfied. It shouldbe noted that a constant resistance per unit length R1′ is a reasonableapproximation in integrated circuits, where conductors are typicallythin compared to skin depth, even at frequencies of several gigahertz.

A single-ended (non-differential) lossless transmission line could beimplemented using a negative distributed transconductance Gm2′.Unfortunately, such an element is not available, so the differentialstructure with cross-coupled control electrodes 14 ⁺ and 14 ⁻effectively emulates a transconductance Gm2′ for the differentialcomponent of the wave in conductors 11 ⁺ and 11 ⁻. Any common-modecomponent will be affected by losses and decay as the wave travels alongthe transmission line.

FIG. 3A depicts one embodiment 30 of a lumped active shunt network. Aninstance of this circuit or its equivalent is placed at regularintervals Δx along the transmission line. In FIG. 1, these intervalswould be at locations 11 a, 11 b, 11 c to 11 n and would be the same onboth differential lines (+ and −). Lumped conductance G2 (14 aG⁺) has avalue G2′·Δx. Similarly, transistors N1 (301) and N2 (302) should have atransconductance Gm2 equal to Gm2′·Δx. This transconductance can beadjusted, for example, by using input terminal bias (304) and transistorN3 (303). Instead of using MOSFETs, as shown, this circuit could as wellbe implemented using other types of transistors or even other circuitelements.

In FIG. 3A, conductors 11 ⁺ and 11 ⁻ are shown on top of conductors 12 ⁺and 12 ⁻ respectively. This configuration is advantageous in integratedcircuit technologies offering a thicker (or wider) metal layer at thetop. The resistance per unit length R1′ of conductors 11 ⁺ and 11 ⁻determines how lossy the passive transmission line would be, hence howmuch power must be spent by the active network in order to compensatefor those losses. For this reason, it is advantageous to allocate thelargest possible cross-section to conductors 11 ⁺ and 11 ⁻. Conversely,there is no particular reason why conductors 12 ⁺ and 12 ⁻ should have alow resistance per unit length, therefore it is acceptable to keep themthin and narrow.

It is well known that the association of a transconductance Gm with aconductance G constitutes an elementary amplifier with a voltage gain Agiven by $\begin{matrix}{A = {- \frac{Gm}{G}}} & (3)\end{matrix}$and an output impedance Zout given by $\begin{matrix}{{Zout} = \frac{1}{G}} & (4)\end{matrix}$

Therefore, the active shunt network shown in FIG. 3A can be replaced byother circuits having the function of an amplifier with the appropriategain and output impedance as shown in FIG. 3B. The distributed nature ofthe ideal active shunt network can be approximated by placing a lumpedamplifier such as 310, 311 at regular intervals Δx along thetransmission line. The interval Δx must be much smaller than theshortest wavelength of interest for the application. This is also trueof the embodiment shown in FIG. 3A. To achieve conditions of losslesspropagation, each amplifier must have a voltage gain $\begin{matrix}{A = {{- \frac{{Gm2}^{\prime}}{{G2}^{\prime}}} = {- \left( {1 + \frac{{C10}^{\prime}}{{C12}^{\prime}}} \right)}}} & (5)\end{matrix}$and an output impedance $\begin{matrix}{{Zout} = {\frac{1}{\Delta\quad{x \cdot {G2}^{\prime}}} = \frac{{M12}^{\prime}}{\Delta\quad{x \cdot {R1}^{\prime} \cdot {C12}^{\prime}}}}} & (6)\end{matrix}$

In this generalized framework, it can be seen that a single-ended (i.e.non-differential) version of the invention can be implemented as well.The schematic of a cross-section of a shunt network for the single-endedtransmission line is shown in FIG. 3C. The main difference is that theamplifier constituting the active shunt network must have a positivegain in this case (non-inverting amplifier). Unfortunately, a positivegain can be obtained in practice only by cascading two amplificationstages with negative gains, which makes the single-ended versionsomewhat less attractive than the differential version. A reasonableimplementation of a single-ended shunt network as shown in FIG. 3D,which is a slightly modified version of the circuit shown in FIG. 3A,yielding a non-inverting amplification stage. The input terminal refmust be set to a constant voltage approximately equal to the DCcomponent of the signal traveling in conductor 11. If one thinks in theframework of amplifiers instead of transconductances, then the benefitof the differential structure is that the amplifiers can be madeinverting (single stage) instead of non-inverting by cross-connectingtheir inputs, whereby a single stage amplifier can be used for eachside.

To avoid reflections at the extremities, each pair of coupledtransmission lines (11 ⁺, 12 ⁺ and 11 ⁻, 12 ⁻) is terminated by thelumped network shown in FIG. 4. At the driven end, a voltage source,such as source 41, can be inserted ahead of conductor 13 and resistanceR0 can be partially or totally absorbed in the output impedance of thevoltage source, if desired.

As will be discussed, reflections will be totally cancelled if theelement values are the following: $\begin{matrix}{{R0} = {v_{p\quad h}{M12}^{\prime}}} & (7) \\{{R1} = {v_{p\quad h}\left( {{L1}^{\prime} - {M12}^{\prime}} \right)}} & (8) \\{{C1} = \frac{1}{v_{p\quad h}{R1}^{\prime}}} & (9)\end{matrix}$

Transmission line parameters defined above in Table 1 apply again inthis case. The phase velocity v_(ph) is equal to the speed of light inthe dielectric material surrounding the transmission line conductors. Itdepends only on the dielectric constant of this material.

The nature and value of impedance Z2 terminating conductor 12 does notaffect propagation conditions in conductor 11. The range of possiblechoices includes the short circuit case (Z2=0), as well as theopen-circuit case (conductor 12 left unconnected).

A 48 mm long transmission line (which could, for example, be used for adelay line, perhaps in a finite impulse response filter) has beensimulated using a circuit simulator. This corresponds to a nominal delayof about 320 ps. The line was modeled by 4800 cascaded segmentsconsisting of lumped capacitors, inductors and resistors. Numericalparameters for this line are listed in Table 2. They have beencalculated using a finite-element approach for a 3.8 μm wide line usingthe top four metal layers of a 0.13 μm CMOS process. TABLE 2 ParameterValue C10′ 173 pF/m C20′ 21.2 pF/m C12′ 97.7 pF/m L1′ 228 nH/m L2′ 518nH/m M12′ 187 nH/m R1′ 2.73 KΩ/m R2′ 109 KΩ/m G2′ 1.43 A/Vm Gm2′ 3.95A/Vm

In a first simulation, the active shunt networks (such as networks 21 ⁺and 21 ⁻ in FIG. 2) were left out. A 1.25 GHz square wave with apeak-to-peak amplitude of 200 mV (differential) was applied to the inputof the termination network. The voltage at nine different taps (51-59)at 6 mm intervals is shown in FIG. 5. As can be seen, edge amplitudedecays rapidly with distance from the driving point.

In a second simulation, shown in FIG. 6, the active shunt networks areadded. Transconductances are modeled by voltage-controlled currentsources. Therefore, the shunt network does not capacitively load theline, and the transconductance value is exact at any frequency. Thewaveforms at the same points (51-59) are shown in FIG. 6. The result isvirtually ideal, free of attenuation and dispersion. This resultconfirms the expectation from theory that a lossless and dispersion-freetransmission line can be achieved within the frequency range where thetransconductances can operate properly. Some moderate ringing, which canbe seen at edges 61, 62, 63, appears to be an artifact of modeling thetransmission line by discrete inductances and capacitances. Such ringingalso occurs in Spice simulations of conventional transmission lines whenlosses are very low and edges very sharp.

In a third simulation, shown in FIG. 7, transconductances areimplemented by actual MOS transistors following the schematic shown inFIG. 3A. The oscillations in 71, 72 and 73 are again ringing, and aremore visible here than in FIG. 6 because the ringing frequency is lowerdue to capacitive loading. Reflections can be seen in the bumps and dipsoccurring after the first 320 ps of the simulation. The waveforms lookalmost perfect for the first 320 ps because the transmission line wasinitially zero at all points. After traveling across the whole line (320ps), the wave hits the termination and is partially reflected backbecause the termination does not perfectly match the transmission linecharacteristic impedance. The reflected wave adds to the forward wavewhich keeps coming from the source. Within ellipses 72 and 73, one canquite distinctly see that the perturbation is also a square wave whichsupports the view that the distortions are due to the addition of thereflected wave.

Some amount of dispersion can also be observed on traces correspondingto taps remote from the driving point. Comparing FIG. 7 to FIG. 6 andlooking at the first 320 ps of simulation, it can be seen that the slopeof the signal becomes gradually less steep as the wave propagates alongthe transmission line. In a perfectly non-dispersive situation, thewaveform would remain identical during propagation. The likely reasonfor this imperfection is that the transconductance of MOSFETs decreasesabove some corner frequency due to channel transit time effects. Abovethis frequency, the conditions for lossless and dispersion-freepropagation may not be able to be fully met.

Instead of only compensating for losses, it is possible to obtain gainfrom a slightly different version. FIG. 8 shows one possiblemodification to achieve network gain. This modification consists ofadding distributed conductance 81 ⁺, 81 ⁻ between conductors 11 ⁺, 11 ⁻and conductor 13, and modifying the value of transconductance Gm2′. Inorder to achieve a specific gain a (measured in Neper per unit length),element values should be chosen as follows: $\begin{matrix}{{G1}^{\prime} = \frac{\alpha^{2}}{{R1}^{\prime}}} & (10) \\{{G2}^{\prime} = \frac{{R1}^{\prime} \cdot {C12}^{\prime}}{{M12}^{\prime}}} & (11) \\{{Gm2}^{\prime} = {{{R1}^{\prime}\frac{{C10}^{\prime} + {C12}^{\prime}}{{M12}^{\prime}}} + \frac{2\alpha}{v_{p\quad h}{M12}^{\prime}} - {\frac{{L1}^{\prime}}{{M12}^{\prime}}\frac{\alpha^{2}}{{R1}^{\prime}}}}} & (12)\end{matrix}$

Unlike conventional distributed amplifiers, there is no fundamentallimit to the length over which this line will provide gain. However,reflections at the terminations would cause the amplifier to oscillateif the total gain multiplied by the reflection coefficient exceededunity. Therefore, the maximum achievable gain is effectively limited bythe accuracy with which reflections can be cancelled (or the reflectionscontained) at the terminations.

FIG. 9 shows generalized shunt network 90 as a theoretical model foridentifying other circuits which can control lossless transmission. Thisshunt network is made of conductances and transconductances betweenconductors 11 ⁺, 12 ⁺, and 13. In FIG. 9, V1 is the voltage betweenconductors 11 ⁺ and 13, and V2 is the voltage between conductor 12 ⁺ and13. As will be discussed, wave propagation on conductor 11 ⁺ will bedispersion-free and present a gain per unit length a if the elementvalues meet the following conditions: $\begin{matrix}{{{Gm1}^{\prime} - {G12}^{\prime} - {Gm12}^{\prime}} = 0} & (13) \\{{{G2}^{\prime} + {G12}^{\prime} + {Gm12}^{\prime}} = \frac{{R1}^{\prime} \cdot {C12}^{\prime}}{{M12}^{\prime}}} & (14) \\{{{- {Gm2}^{\prime}} + {G12}^{\prime} + {Gm21}^{\prime}} \geq {{\frac{{L1}^{\prime}}{{M12}^{\prime}}\frac{\alpha^{2}}{{R1}^{\prime}}} + {{R1}^{\prime} \cdot \left( {{C10}^{\prime} + {C12}^{\prime}} \right)}}} & (15) \\{{{G1}^{\prime} + {G12}^{\prime} + {Gm21}^{\prime}} = {\frac{{M12}^{\prime}}{{L1}^{\prime}}\left( {{- {Gm2}^{\prime}} + {G12}^{\prime} + {Gm21}^{\prime} - {{R1}^{\prime} \cdot \left( {{C10}^{\prime} + {C12}^{\prime}} \right)}} \right)}} & (16)\end{matrix}$

There are seven free parameters and only four constraints, thereforethere are potentially many choices. The solution described with respectto FIG. 1 corresponds to the case where Gm1′, G12′, Gm12′, Gm21′ and G1′are all set to zero. This is believed to be the simplest circuit meetingthese constraints, but other solutions exist.

It is not possible to represent all possible circuits meeting theseconstraints in a single schematic because the connectivity of thenetwork depends on the signs of element values. Negative element valuescan be implemented, for example, by cross-connecting the element betweenthe two sides of a differential structure, as shown in FIGS. 1 and 2,whereas positive parameters correspond to elements connected solely toone side of the structure. It would be a tedious job to draw allpossible solutions. Instead, just a single alternative solution is shownin FIG. 10 for the purpose of illustrating how diverse other solutionscan become. This circuit corresponds to G12′=0, Gm12′=0 and Gm2′=0,which leads to a positive Gm21′, a positive G2′ and a negative G1′. Thiscircuit has practical drawbacks compared to the preferredimplementation, but in principle it would also work.

The distributed structure described in FIG. 1 can be approximated by acircuit made entirely of lumped elements. This lumped approximation isuseful if the purpose of the circuit is to delay a signal by a specificamount, rather than transmitting the signal from one point to another.In this case, the coupled transmission lines are replaced by a laddernetwork of coupled lumped inductors and capacitors. A discrete activeshunt network circuit is added at each node of the ladder network inorder to cancel resistive losses in the inductors.

A schematic of such an embodiment is shown in FIG. 11. The dashed arrowsindicate that the controlled current sources Gm2 ⁺ are controlled by thevoltage on conductor 11 ⁻ on the opposite side of the differentialstructure. Two stages of a single-ended network are shown, but thenetwork may have any number of such stages in cascade. Only one side(the positive side) of the differential structure is shown. The second(negative) side is identical. Only intentional circuit elements areshown. The inductors unavoidably have some parasitic resistance inseries which causes losses. The active shunt networks consisting of aconductance G2 and a controlled current source Gm2 cancel these losses.In practice, the active shunt networks can be implemented by the samecircuit as shown in FIG. 3.

It can be shown that network 1100 constitutes a delay line. The signalapplied to conductor 11 at one end of the line propagates at a constantvelocity along the network. Each stage introduces a delay Δt given by$\begin{matrix}{{\Delta\quad t} = \sqrt{{L1} \cdot \left( {{C10} + \frac{{C12} \cdot {C20}}{{C12} + {C20}}} \right)}} & (17)\end{matrix}$

Therefore, each stage approximates a length v_(ph)·Δt of distributedtransmission line, where v_(ph) is the velocity of light in the mediumunder consideration.

In order to achieve lossless and dispersion-free propagation, elementvalues shown in FIG. 11 must be related as specified in the followingequations: $\begin{matrix}{\quad{{G2} = \frac{{R1} \cdot {C12}}{M12}}} & (18) \\{{Gm2} = {{R1}\frac{{C10} + {C12}}{M12}}} & (19) \\{{M12} = {\frac{C12}{{C12} + {C20}}{L1}}} & (20)\end{matrix}$

A number of possible applications of this concept exist. One such is inthe transmission of high-speed signals across a large chip. Another isan amplifier. Two other applications will be described with respect toFIGS. 12A and 12B.

A finite impulse-response filter computes a discrete weighted sum ofdelayed copies of its input signal: $\begin{matrix}{{Y(t)} = {\sum\limits_{k = 1}^{N}\quad{W_{k} \cdot {x\left( {t - {\Delta\quad t_{k}}} \right)}}}} & (21)\end{matrix}$

The operation performed by the finite impulse response filter depends onthe values of coefficients W_(k), known as tap weights. The delaysΔt_(k) are typically integer multiples of some unit delay. Two possibleanalog implementations of a finite impulse response filter areillustrated in FIGS. 12A and 12B.

FIG. 12A shows the input signal 1201 applied to delay line 1202. Tapsalong this line provide delayed copies of the input signal. The voltageat each tap is applied to a transconductance amplifier such as 1203-1delivering a current proportional to the voltage. The ratio Gmk betweencurrent and voltage determines the tap weight. The currents from alltransconductance amplifiers are added together on a global node output1204 loaded by resistor R_(load). The voltage on this global node is theoutput signal of the filter. This circuit works in principle, but it maybe difficult in practice to reach very high speeds because the parasiticcapacitance of the global output node tends to be large.

A slightly modified architecture is shown in FIG. 12B. In thisvariation, delay elements 1205 are added between the outputs of thetransconductance amplifiers, whereby the delays on the input side arecorrespondingly reduced in order to maintain the same total delay as thesignal travels from the input to the output through a given tap. Thiscircuit operates in much the same way as does the circuit shown in FIG.12A, but can achieve much higher bandwidth because the tap outputs areaggregated over a delay line. The reason is that the parasiticcapacitance unavoidably present at the output of each transconductanceamplifier becomes part of the delay line capacitance in this case. Animplementation of this architecture using passive (i.e. lossy)inductor/capacitor ladders as delay lines has been published recently(see e.g., Wu, et al., “Differential 4-Tap and 7-Tap Transverse Filtersin SiGe for 10 Gb/s Multimode Fiber Optic Link Equalization”, IEEE Int.Solid-State Circ. Conf., San Francisco, February 2003, hereinafterreferred to as Wu). The Wu filters could be implemented using thetechniques shown in FIGS. 12A and 12B.

Both architectures could be implemented either in the form of adistributed transmission line, or in the form of a lumped approximation.

As discussed above, the transmission line concepts can be used as anamplifier. If both ends of the transmission line are terminated asdescribed with respect to FIG. 4, then a signal applied to one end willtravel only one way across the line, and will be fully absorbed by thetermination. If the termination does not perfectly match thecharacteristic impedance of the line, then at least a fraction of thesignal will be reflected back and amplified again on the return to thesource. Again, if the source impedance does not match the characteristicimpedance of the line, a fraction of the signal is reflected back intothe line.

If the total gain of the amplifying line is A, and if a fraction α ofthe signal is reflected back at each termination, then the line willbecome unstable if the product α·A exceeds unity. In this case, anoscillation will build up from noise and the line can be used as anoscillator. The fundamental oscillation frequency f₀ will be$\begin{matrix}{f_{0} = \frac{v_{ph}}{2L}} & (22)\end{matrix}$where v_(ph) is the speed of light in the considered medium and L is thelength of the line. The oscillator output will generally also includeharmonics of this frequency.

An economical way to produce such an oscillation would be to leave bothends of the line open (no termination), whereby α would approach unity.In this case, gain just slightly larger than unity should suffice toproduce sustained oscillations.

In an integrated circuit, both v_(ph) and L can be accurately controlledand stable over time and environmental conditions, therefore it shouldbe possible to use such an oscillator as an on-chip frequency reference.

Wave propagation characteristics of a pair of coupled lines with adistributed active shunt network will now be derived. FIG. 13 (shown onthe page with FIG. 4) shows equivalent circuit 1300 of a short segmentof the structure under consideration. Distributed capacitances C110′,C20′ and C12′, inductances L1′, L2′, M12′ and series resistances R1′ andR2′ are inherent to the transmission line conductors. Shunt conductancesG1′, G2′ and transconductance Gm2′ are added in an attempt to make theline dispersion-free and lossless. The ground conductor 13 carrying thereturn currents is not explicitly shown.

It should be noted that the set of transmission line parameters isredundant if propagation is purely transverse electro-magnetic (TEM).Inductance parameters L1′, L2′ and M12′ are related to capacitanceparameters C10′, C20′ and C12′. It can be shown that the relationshipbetween them is: $\begin{matrix}{{L1}^{\prime} = {\frac{{C12}^{\prime} + {C20}^{\prime}}{{{C10}^{\prime}{C20}^{\prime}} + {{C10}^{\prime}{C12}^{\prime}} + {{C20}^{\prime}{C12}^{\prime}}} \cdot \frac{ɛ_{r}}{c_{0}^{2}}}} & (23) \\{{L2}^{\prime} = {\frac{{C12}^{\prime} + {C10}^{\prime}}{{{C10}^{\prime}{C20}^{\prime}} + {{C10}^{\prime}{C12}^{\prime}} + {{C20}^{\prime}{C12}^{\prime}}} \cdot \frac{ɛ_{r}}{c_{0}^{2}}}} & (24) \\{{M12}^{\prime} = {\frac{{C12}^{\prime}}{{{C10}^{\prime}{C20}^{\prime}} + {{C10}^{\prime}{C12}^{\prime}} + {{C20}^{\prime}{C12}^{\prime}}} \cdot \frac{ɛ_{r}}{c_{0}^{2}}}} & (25)\end{matrix}$

In these equations, c₀ is the speed of light in vacuum and ε, is thedielectric constant of the medium surrounding the conductors.

The voltage gradient at a given point of the transmission line isrelated to the currents at this point as follows: $\begin{matrix}\left\{ \begin{matrix}{\frac{\partial V_{1}}{\partial x} = {{{- \left( {{R1}^{\prime} + {sL1}^{\prime}} \right)} \cdot I_{1}} - {{sM12}^{\prime} \cdot I_{2}}}} \\{\frac{\partial V_{2}}{\partial x} = {{{- {sM12}^{\prime}} \cdot I_{1}} - {\left( {{R2}^{\prime} + {sL2}^{\prime}} \right) \cdot I_{2}}}}\end{matrix} \right. & (26)\end{matrix}$

Similarly, the current gradient can be written: $\begin{matrix}\left\{ \begin{matrix}{\frac{\partial I_{1}}{\partial x} = {{{- \left( {{G1}^{\prime} + {s\left( {{C10}^{\prime} + {C12}^{\prime}} \right)}} \right)} \cdot V_{1}} + {{sC12}^{\prime} \cdot V_{2}}}} \\{\frac{\partial I_{2}}{\partial x} = {{{- \left( {{- {Gm2}^{\prime}} + {sC12}^{\prime}} \right)} \cdot V_{1}} - {\left( {{G2}^{\prime} + {s\left( {{C20}^{\prime} + {C12}^{\prime}} \right)}} \right) \cdot V_{2}}}}\end{matrix} \right. & (27)\end{matrix}$

It is helpful to rewrite these equations in matrix form: $\begin{matrix}\left\{ {\begin{matrix}{{\frac{\partial\quad}{\partial x}\begin{pmatrix}V_{1} \\V_{2}\end{pmatrix}} = {{- Z} \cdot \begin{pmatrix}I_{1} \\I_{2}\end{pmatrix}}} \\{{\frac{\partial\quad}{\partial x}\begin{pmatrix}I_{1} \\I_{2}\end{pmatrix}} = {{- Y} \cdot \begin{pmatrix}V_{1} \\V_{2}\end{pmatrix}}}\end{matrix}{where}} \right. & (28) \\{Z = \begin{pmatrix}{{R1}^{\prime} + {sL1}^{\prime}} & {sM12}^{\prime} \\{sM12}^{\prime} & {{R2}^{\prime} + {sL2}^{\prime}}\end{pmatrix}} & (29) \\{Y = \begin{pmatrix}{{G1}^{\prime} + {s \cdot \left( {{C10}^{\prime} + {C12}^{\prime}} \right)}} & {- {sC12}^{\prime}} \\{{Gm2}^{\prime} - {sC12}^{\prime}} & {{G2}^{\prime} + {s \cdot \left( {{C20}^{\prime} + {C12}^{\prime}} \right)}}\end{pmatrix}} & (30)\end{matrix}$

Currents can be eliminated from equation (28) in order to get asecond-order differential equation of voltages: $\begin{matrix}{{{\frac{\partial^{2}}{\partial x^{2}}\begin{pmatrix}V_{1} \\V_{2}\end{pmatrix}} = {- {\gamma^{2}\begin{pmatrix}V_{1} \\V_{2}\end{pmatrix}}}}{where}} & (31) \\{\gamma^{2} = {Z \cdot Y}} & (32)\end{matrix}$

In order to achieve dispersion-free wave propagation in conductor 11,the matrix γ² must have the following form: $\begin{matrix}{\gamma^{2} = \begin{pmatrix}\left( {{- \alpha} + \frac{s}{v_{ph}}} \right)^{2} & 0 \\\gamma_{2l}^{2} & \gamma_{22}^{2}\end{pmatrix}} & (33)\end{matrix}$the gain exponent α must be real and positive or null. The quantityv_(ph) is the phase velocity of the wave in conductor 11. Assuming thata matrix γ² of this form can be achieved, the solution of equation (31),as far as conductor 11 is concerned, can be written:V ₁(x)=A _(f)·exp(−γ₁ x)+A _(r)·exp(γ,x)  (34)

Parameters A_(f) and A_(r) are constants which must be determined usingboundary condition at both ends of the transmission line. Thepropagation exponent is $\begin{matrix}{\gamma_{1} = {{- \alpha} + \frac{s}{v_{ph}}}} & (35)\end{matrix}$

The form of this propagation exponent implies that the magnitude of thewave traveling across conductor 11 grows exponentially with distance andhas linear phase, hence no dispersion.

It remains to be shown that an appropriate choice of G1′, G2′ and Gm2′will indeed produce an exponent matrix γ² of the desired form. For thispurpose, the first row of the matrix product ZY must be written out indetail using equations (29) and (30) and made equal to the first row ofmatrix γ² defined in equation (33). The resulting set of equations mustbe solved for G1′, G2′ and Gm2′. This simple but somewhat tediousprocedure leads to the result that γ² has the required form if thefollowing relations are satisfied: $\begin{matrix}{{G1}^{\prime} = \frac{\alpha^{2}}{{R1}^{\prime}}} & (36) \\{{G2}^{\prime} = \frac{{R1}^{\prime}{C12}^{\prime}}{M12}} & (37) \\{{- {Gm2}^{\prime}} = {{{R1}^{\prime}\frac{{C10}^{\prime} + {C12}^{\prime}}{{M12}^{\prime}}} - \frac{2\alpha}{v_{ph}{M12}^{\prime}} + {\frac{{L1}^{\prime}}{{M1}^{\prime}}\frac{\alpha^{2}}{{R1}^{\prime}}}}} & (38)\end{matrix}$

As one could have expected, the phase velocity v_(ph) of the wavetraveling across conductor 11 turns out to be: $\begin{matrix}{v_{p\quad h} = \frac{c_{0}}{\sqrt{ɛ_{r}}}} & (39)\end{matrix}$

Equations (36)-(38) were already introduced above as equations withoutdemonstration.

If the generalized active shunt network from FIG. 9 is consideredinstead of the simpler version of FIG. 13, matrix Y will be somewhatdifferent, but the same procedure can be applied to find the resultpresented above.

The characteristic impedance of a pair of coupled lines cannot generallybe expressed by a single scalar, but rather by a matrix Z_(c). Thismatrix is the ratio between the series impedance matrix Z defined inequation (29) and the propagation exponent of the wave traveling in theline.

In the lossless case, α=0, and the characteristic impedance matrix hasthe following form: $\begin{matrix}{Z_{c} = \begin{pmatrix}{\frac{v_{p\quad h}{R1}^{\prime}}{s} + {v_{p\quad h}{L1}^{\prime}}} & {v_{p\quad h}{M12}^{\prime}} \\Z_{c21} & Z_{c22}\end{pmatrix}} & (40)\end{matrix}$

The terms Z_(c21) and Z_(c22) can be calculated but the resultingexpressions are rather intricate and of little relevance to signalpropagation on conductor 11. For this reason, the detailed expressionsare not provided here.

A line of finite length must be terminated by a lumped networkcharacterized by the same impedance matrix as Z_(c). The terminationnetwork shown in FIG. 4 meets this constraint. If some finite gain orloss is present (α≠0), the characteristic impedance matrix becomes morecomplicated and any termination network that would match the impedanceof the coupled line perfectly is not a simple matter. Thus, α should beas small as possible to achieve decent results.

The disclosed transmission line is characterized by the propagationexponent shown in equation (35), which corresponds to the form requiredto achieve dispersion-free propagation. Equation (35) is written in theLaplace domain and therefore uses the Laplace variable s which is equalto jw. A negative sign for α is used in equation (35) because thecircuit achieves gain. In the more conventional form (positive sign infront of α), a positive value of α means that there are losses, whereasa negative value means that there is gain. Knowing that the inventionhas gain, it is best to use the opposite convention so that positivevalues of α mean gain.

Note that the concepts discussed herein become relevant at frequencieswhere the length of the transmission line reaches or exceeds about ¼wavelength. With a signal of 1 GHz in a medium where electromagneticwaves propagate at about half the speed of light in vacuum, a wirestarts to look like a transmission line if its length exceeds about 37.5mm. Losses become important at this point. The value of 1 GHz isreasonable for use inside integrated circuits or on printed circuitboards. If long-distance, propagation is considered (on the order ofmeters or even kilometers), then transmission line theory applies atmuch lower frequencies, and losses would have to be considered as well.

Although the present invention and its advantages have been described indetail, it should be understood that various changes, substitutions andalterations can be made herein without departing from the invention asdefined by the appended claims. Moreover, the present application is notintended to be limited to the particular embodiments of the process,machine, manufacture, composition of matter, means, methods and stepsdescribed in the specification. As one will readily appreciate from thedisclosure, processes, machines, manufacture, compositions of matter,means, methods, or steps, presently existing or later to be developedthat perform substantially the same function or achieve substantiallythe same result as the corresponding embodiments described herein may beutilized. Accordingly, the appended claims are intended to includewithin their scope such processes, machines, manufacture, compositionsof matter, means, methods, or steps.

1. An amplifier comprising: a first primary conductor; a first auxiliaryconductor inductively coupled to said first primary conductor; andnon-inverting amplification with an input connected to said primaryconductor and an output connected to said first auxiliary conductor,said amplification being distributed along the length of said primaryconductor and adjusted to provide overall gain along said primaryconductor.
 2. The amplifier of claim 1 further comprising: a groundconductor; and wherein said non-inverting amplification comprises: afirst transistor having its gate connected to said first primaryconductor, its drain connected to said ground conductor, and its sourceconnected to the source of a second transistor; and said secondtransistor having its gate connected to a reference voltage input, itsdrain connected to said first auxiliary conductor and through aconductance to said ground conductor.
 3. The amplifier of claim 2further comprising: a current sink connected to said sources of saidfirst and second transistor.
 4. The amplifier of claim 1 wherein saidnon-inverting amplification comprises: non-inverting amplifiers atspaced intervals along said transmission line.
 5. The amplifier of claim4 wherein said spaced intervals are equal.
 6. An oscillator comprising:an amplifier in accordance with claim 1, and further comprising: areflective termination for returning a substantial portion of a signalpassing along said primary conductor.
 7. The oscillator of claim 6wherein the fraction of the signal reflected by said reflectivetermination when multiplied by said overall gain is greater than unity.8. A differential amplifier comprising: first and second primaryconductors; a first auxiliary conductor inductively coupled to saidfirst primary conductor; a second auxiliary conductor inductivelycoupled to said second primary conductor; first inverting amplificationwith an input connected to said first primary conductor and an outputconnected to said second auxiliary conductor; and second invertingamplification with an input connected to said second primary conductorand an output connected to said first auxiliary conductor, said firstamplification and second amplification distributed along saidtransmission line, said first amplification and said secondamplification providing overall gain along said first and second primaryconductors.
 9. The amplifier of claim 8 further comprising: a groundconductor common to said primary conductors and common to said auxiliaryconductors; and wherein each said first amplification and said secondamplification has conductance with respect to said ground conductor. 10.The amplifier of claim 8 wherein said first amplification and saidsecond amplification comprise inverting amplifiers spaced along saidprimary conductors.
 11. The amplifier of claim 10 wherein each saidamplifier comprises: a transistor and wherein the source of saidtransistors are connected together and also connected to a bias input.12. The amplifier of claim 11 wherein said bias input comprises: acurrent source.
 13. An oscillator comprising an amplifier in accordancewith claim 8 and further comprising: reflective terminations reflectinga fraction of an incident signal applied to an input of said amplifier,said fraction such that the amplitude of the reflected signal multipliedby the amplifier gain exceeds the amplitude of said incident signal. 14.An oscillator comprising an amplifier in accordance with claim 8 andfurther comprising: a reflective termination for returning a substantialportion of a signal passing along said primary conductor.
 15. Theoscillator of claim 14 wherein the fraction of the signals reflected bysaid reflective termination when multiplied by said overall gain isgreater than unity.
 16. A method for transporting a.c. signal to achieveoscillation, said method comprising: propagating the a.c. signal along aprimary conductor; coupling said propagating signal from said primaryconductor to an auxiliary conductor along the length of said primaryconductor; establishing both conductance and transconductance betweensaid auxiliary conductor and a ground conductor; establishing an overallgain along said conductors; and reflecting said signals back at thetermination of said conductors to achieve oscillation.
 17. The method ofclaim 16 wherein reflected signals when multiplied by said overall gainare greater than unity.
 18. The method of claim 16 wherein saidtransconductance is negative.
 19. The method of claim 16 wherein saidcoupling comprises: inductively coupling said a.c. signal.
 20. Themethod of claim 19 wherein said inductively coupling comprises:inductively coupling said propagating a.c. signal from said primaryconductor to said auxiliary conductor along the length of said primaryand auxiliary conductors.